FIG. 1 and FIG. 2 show an exemplary configuration of the spreading, modulation, and framing design for the uplink (i.e. the mobile station (MS) to base station (BS) link) of a so-called "3.sup.rd Generation" (3G) cellular communications system based on Code Division Multiple Access (CDMA) principles. That system is generally described in the European Telecommunications Standards Institute (ETSI) report of sub-group SMG2, entitled "Concept Group Alpha--Direct-Sequence CDMA (WCDMA) Evaluation Document (3.0), Part1, Dec. 15-19, 1997." There are two types of physical sub-channels shown in FIG. 1, namely the Dedicated Physical Data Channel (DPDCH) 101 and the Dedicated Physical Control Channel (DPCCH) 102. The DPDCH is used to carry dedicated data generated at layer 2 and above while the DPCCH is used to carry control information generated at layer 1.
As shown in FIG. 2, the DPDCH and DPCCH are separated into 10 ms frames 201, and further separated into 625 us timeslots 202. The DPDCH carries 10*2.sup.K BPSK symbols (i.e. bits) in each timeslot, where K is an integer in the range 0-6. In the example illustrated by FIG. 2, K=1, allowing 20 data bits 206 (symbols d(0)-d(19)) to be transmitted per timeslot giving a bit rate of 32 kbps. Since the spreading rate is 4.096 Mega-chips per second in the example shown, there are 2560 (N.sub.P) chips in each timeslot, with each DPDCH data symbol being spread by a factor of 128.
The DPCCH consists of a) known pilot symbols 203 to support channel and received signal-to-noise (SNR) estimation at the BS rake receiver, b) a Transmit Power Control Indicator (TPCI) field 204 to support closed-loop BS transmit power control, and c) a Rate Indicator (RI) field 205 used to describe the information rate of each 10 ms frame. The DPCCH is typically spread by a factor of 256.
The DPDCH and DPCCH are respectively mapped to the in-phase (I) and quadrature (Q) branches of the MS transmitted waveform. The resulting I and Q signal is then spread to the chip rate using two different orthogonal channelization codes C.sub.D 103 and C.sub.C 104. The channelization codes are Orthogonal Variable Spreading Factor (OVSF) codes selected from the hierarchy of such codes described in detail in "Tree Structured Generation of Orthogonal Spreading Codes with different lengths for Forward Link of DS-CDMA Mobile Radio," Electronics Letters, Jan. 2nd, 1997, pp.27-28, by F. Adachi, M. Sawahashi, and K. Okawa. Since, in the example shown, it is assumed that the desired coded bit rate of the DPCCH is 32 kbps, the DPDCH channelization code C.sub.D is required to be length 128. The DPCCH channelization code is likewise an allowable OVSF code of length 256, regardless of the bit rate of the DPDCH.
The complex-valued signal 111 resulting from the combination of DPCCH and DPDCH is then scrambled by an MS-specific primary scrambling code C.sub.S 105. In the prior art, this primary scrambling code is complex-valued, is of length-256, and is constructed using a pair of extended Very Large Kasami codes although any code family with good auto- and cross-correlation properties may be used. The resulting spread signal is then filtered using square-root raised-cosine chip pulse-shaping filters 108 and 109 before being quadrature modulated 109 and transmitted from antenna 110.
Radio trilateration techniques have been widely used in many mobile transmitter location systems (e.g., Loran-C). There are two primary methods to achieve trilateration, namely the a) Time of Arrival (TOA) and b) Time Difference of Arrival (TDOA) methods. TDOA requires each BS to estimate, with respect to a local timing reference, the time delay of the MS signal using a time delay estimator. FIG. 3 shows the block diagram of a prior-art BS time delay estimator that may be used to form such an estimate given the transmitted MS signal described above. In FIG. 3, the received signal is first converted to baseband using quadrature demodulator 301, then processed using chip pulse matched filter 302, and sampled using analog-digital (ADC) 303. The signal is then passed through Code Matched Filter (CMF) 304, where the taps of the filter are set equal to the length-N.sub.PILOT code C* 207 shown in FIG. 2, where C* 207 is constructed by concatenating K copies of the length-256 code formed as the product of the C.sub.C and C.sub.S codes, and where N.sub.PILOT =256K. In the example of FIG. 3, K=5.
A time delay estimate is then formed by coherently accumulating 305 the complex-valued CMF outputs to an array of N.sub.P observations or "ranging bins" (where N.sub.P is 2560 as shown in FIG. 2) for N timeslots, and then non-coherently accumulating 312 the result for M such length-N timeslot periods. The total observation time is therefore N.times.M timeslots. The length-N.sub.P real-valued output array of the non-coherent accumulator 312, i.e., the array of available decision statistics 307, is then passed through a Time Delay Decision Algorithm 308 which gives the time delay estimate to 309 of the MS signal with respect to the local timing reference, where the Time Delay Decision Algorithm 308 may be as simple as selecting the ranging bin with largest magnitude decision statistic 307. This time delay estimate 309 can then be combined with time-delay estimates from other BSs and used in the derivation of MS location using, for example, the TDOA technique.
The ability to use such trilateration techniques in a DS-CDMA system utilizing MS transmit power control is, however, problematic since MSs close to BSs (i.e. a "proximal" BS) transmit at low output power in order to avoid the well-known near-far problem for CDMA systems. Since such MSs are observed at very low SNR at distant BSs, it may not be possible in such cases to form a reliable time-delay estimate at distant BSs. One of the ways to increase the visibility of the MS to multiple BSs is to increase the MS power level for a brief period at a predefined time period using a so-called Power-Up Function (PUF) command from the infrastructure equipment. Details of PUF can be found in Telecommunications Industry Association (TIA) contribution "Location Power Up Function," TR45.5.2.3/97.07.17.02 by A. Ghosh, G. Bruckert, B. Verbiscer and M. Panjwani and in ETSI SMG2 document UMTS A36/97 and U.S. Pat. No. 5,508,708 dated Apr. 16, 1996 by Ghosh et. al.
PUF can be used for performing MS location in systems employing the prior-art MS transmitter/transmission structure described above. In this case, the pilot field of the DPCCH (see FIG. 1 and FIG. 2) is sent at higher power for the duration of the PUF. The TDOA information required for MS location can then be derived from observation of the pilot field at multiple distant base stations using trilateration methods and a code matched filter (CMF) matched only to the pilot field of the DPCCH. (If the entire DPCCH transmission over the timeslot were used, range estimation would have to be performed jointly with determination of modulated TPCI 204 and RI 205 symbols). In practice, however, the CMF would not be matched to the entire DPCCH and would not be of length-2560, but rather of length N.sub.PILOT. This leads to reduction in SNR at distant BS filter outputs. Also, it is inefficient to transmit the TPCI 204 and RI 205 fields at higher power during PUF, since those fields are not used at the CMF. Transmitting only the pilot field 203 at higher power, however, would lead to undesirable MS transmit power envelope variations. Furthermore PUF probe cancellation at proximal BSs becomes cumbersome with this approach.
Thus a need exists for a method and apparatus for location of a mobile station within a communication system that results in a higher SNR at distant BSs without the undesirable MS transmit power envelope variations and other disadvantages of prior-art location techniques.